This project,a stepper
motor emf position detector from 1992,
It works by detecting the emf generated by the coils as they go past each
magnet when the coils are powered down (as they will be during each part of the
stepper motor driving cycle).
A method for measuring the position of a stepping motor by generating a
signal representative of the back electromotive force induced by the rotor in
the coils of the motor, and estimating the actual position of the rotor from
the signal representative of the back electromotive force induced by the
rotor.
The present designs relates generally to the determination of the angular
position of a rotating apparatus, and more specifically to the accurate
determination of the position of a D.C. stepping motor.
DESCRIPTION OF THE PRIOR ART
In the measurement of position or velocity of rotating elements, such as
drivers for a xerographic photoreceptor belt, it is common practice to utilize
encoders to monitor the position of the rotating element. In general, however,
the accuracy of the positional information is limited to the resolution of the
encoder used.
It is also generally known that in order to increase the accuracy of the
positional information, more expensive, high resolution encoders must be
employed.
It is known that closed loop control of a motor requires that information
regarding the shaft position, that is, velocity or position, be available to
the controller. For many motors, this information is provided by an external
sensor, such as a tachometer or incremental position encoder. This external
sensor represents a substantial additional cost to the motion control
system.
In the field of motion control, it is common practice to use stepping motors,
operating in open-loop fashion, i.e., without the use of position and/or
velocity feedback devices, as a cost effective alternative to closed loop
operation of DC motors with their attendant and necessary feedback elements,
such as tachometers and/or incremental position encoders.
While widely practiced, it is just as widely known that there are two
undesirable characteristics of open-loop stepping motor operation that must be
considered and/or overcome in the design of a successful motion control
application: (1) stalling of the stepping motor during load torque or speed
changes; (2) the highly underdamped, resonant response commonly referred to as
"ringing".
As seen by a review of the prior art, great progress has been made in
preventing the occurrence of stalling during open-loop stepper operation by
properly timing the occurrence of step commands with knowledge of motor
position gleaned from examination of the back EMF of the stepping motor coils,
a process commonly called self-synchronization.
In the present design, methods are developed to yield position estimates of
sufficient resolution and accuracy throughout the entire electrical cycle to
enable true closed loop control of stepper motor operation. In this way the
second major defect of open loop stepper operation, i.e., "ringing" can be
controlled.
As is well know by those skillful in the art, closed loop feedback control
allows the designer of a dynamic system the opportunity to change the closed
loop system's poles or eigenvalues and thus to shape the closed loop response
of the system. Specifically with respect to stepping motors, true closed loop
feedback control means the opportunity to eliminate "ringing" by shaping the
closed loop system's dominant pair of oscillatory eigenvalues to be well
damped.
A major difference between the prior art and the present design is that in the
prior art electrical cycle position estimates are generated only at a few
selected points in the electrical cycle, e.g., 0, 90, 180,270 degrees, for the
purpose of self-synchronizing full stepping open loop stepper operation.
The present design has the capability of making positional measurements
throughout the electrical cycle at any time without using op-amp coil
simulation, or specialized sensing coils. In the prior art, the measurement of
the back EMF is often done by either adding voltage sensing coils to the
stepper motor and/or simulating the coil via analog op-amp circuitry.
The need for sensing coils limits closed loop control of stepping motors to
specially constructed motors which are more expensive than the common stepper
motor used ill many motion control applications and thus excludes the vast
majority of motors commonly available. The op-amp approach, as seen, for
example, in U.S. Pat.
No. 4,658,194 to Richter et al., has other disadvantages that are impractical.
In particular, it uses the simulation of a differentiator to estimate the
voltage drop across the inductor of the motor coil. As well know to anyone
skillful in the art, differentiation is an inherently noisy process and thus a
serious signal to noise problem is introduced in the back EMF measurement which
in turn limits the positional accuracy of the estimate.
Also, the op-amp approach adds additional circuitry to the system, and thus,
additional cost.
It would be desirable, therefore, to provide true closed loop control of the
motor itself rather than merely self synchronization. It would also be
desirable to provide closed loop control of a hybrid stepper motor allowing the
stepper to be used in critical constant velocity systems, such as those in a
xerographic reproduction machine.
It is an object, therefore, of the present design, to provide a method to
determine the position of a hybrid stepping motor throughout the entire
electrical cycle, allowing accurate position measurement of the stepper motor
without an external sensor. It is another object of the present design to
provide a self encoding technique incurring virtually no additional cost over
an open loop stepper motor system while reaping the benefits of closed loop
control.
It is still another object of the present design to provide a self encoding
technique that uses the fact that a relationship exists between the back EMF of
the motor coils and the electrical position and virtually no additional
hardware is needed for sensing the coil back EMF.
A method for measuring the position of a stepping motor having a rotor and a
plurality of coils including generating a signal representative of the back
electromotive force induced by the rotor in the coils, and estimating the
actual position of the rotor from the signal representative of the back
electromotive force induced by the rotor.
View larger image here.
For a general understanding of the features of the present design, reference
is made to the drawings. Referring to FIG. 1, there is shown a typical
xerographic reproduction machine 5 composed of a plurality of programmable
components and subsystems which cooperate to carry out a copying or printing
job.
The machine employs a photo-conductive belt 10, which is entrained about
stripper roller 14, tensioning roller 16, idler rollers 18, and drive roller
20. Drive roller 20 is rotated by a conventional motor (not shown) coupled
thereto by suitable means such as a belt drive. Drive roller 20 is also
operatively connected to a shaft encoder having a resolution of 100
lines/revolution, whereby the belt position and velocity are tracked via
signals from the encoder As roller 20 rotates it advances belt 10 in the
direction of arrow 12 through the various processing stations disposed about
the path of movement thereof.
Initially, the photoconductive surface of belt 10 passes through charging
station A where corona generating devices, indicated generally by the reference
numerals 22 and 24, charge photoconductive belt 10 to a relatively high,
substantially uniform potential. Next, the charged photoconductive belt is
advanced through imaging station B.
At imaging station B, a document handling unit 26 sequentially feeds documents
from a stack of documents in document tray 27 into registered position on
platen 28. Xenon flash lamps 30 mounted in optics cavity 31 illuminate the
document on platen 28, the light rays reflected from the document being
focussed by lens 32 onto belt 10 to expose an electrostatic latent image on
photoconductive belt 10 which corresponds to the informational areas contained
within the document currently registered on platen 28.
After imaging, the document is returned to document tray 27 via a simplex copy
path or if the first pass of a duplex copy is being made via a duplex path.
The electrostatic latent image recorded on photoconductive belt 10 is developed
at development station C by a magnetic brush developer unit 34 having developer
roll assemblies 36, 38 and 40. A paddle wheel 42 picks up developer material
and delivers it to the developer roll assemblies 36, 38. Developer roll
assembly 40 is a cleanup roll while magnetic roll 44 is provided to remove any
carrier granules adhering to belt 10.
Following development, the developed image is transferred at transfer station D
to a copy sheet provided via de-skew rollers 71 and paper feed roller 72. Tile
re, the photoconductive belt 10 is exposed to a pre-transfer light from a lamp
(not shown) to reduce the attraction between photoconductive belt 10 and the
toner powder image.
Next, a corona generating device 46 charges the copy sheet to the proper
magnitude and polarity so that the copy sheet is tacked to photoconductive belt
10 and the toner powder image attracted from the photoconductive belt to the
copy sheet. After transfer, corona generator 48 charges the copy sheet to the
opposite polarity to detack the copy sheet from belt 10.
Following transfer, a conveyor 50 advances the copy sheet bearing the
transferred image to fusing station E where a fuser assembly, indicated
generally by the reference numeral 52 permanently affixes the toner powder
image to the copy sheet. Preferably, fuser assembly 52 includes a heated fuser
roller 54 and a pressure roller 56 with the powder image on the copy sheet
contacting fuser roller 54.
After fusing, the copy sheets are fed through a decurler 58 to remove any curl.
Forwarding rollers 60 then advance the sheet via duplex turn roll 62 to gate 64
which guides the sheet to either finishing station F or to duplex tray 66, the
latter providing an intermediate or buffer storage for those sheets that have
been printed on one side and on which an image will be subsequently printed on
the second, opposed side thereof. The sheets are stacked in duplex tray 66 face
down on top of one another in the order in which they are copied.
To complete duplex copying, the simplex sheets in tray 66 are fed, in seriatim,
by bottom feeder 68 back to transfer station D via conveyor 70, de-skew rollers
71 and paper feed rollers 72 for transfer of the second toner powder image to
the opposed sides of the copy sheets. The duplex sheet is then fed through the
same path as the simplex sheet to be advanced to finishing station F.
Copy sheets are supplied from a secondary tray 74 by sheet feeder 76 or from
auxiliary tray 78 by sheet feeder 80. Sheet feeders 76, 80 are friction retard
feeders utilizing a feed belt and take-away rolls to advance successive copy
sheets to transport 70 which advances the sheets to rolls 72 and then to
transfer station D.
A high capacity feeder 82 is the primary source of copy sheets. Tray 84 of
feeder 82, which is supported on an elevator 86 for up and down movement, has a
vacuum feed belt 88 to feed successive uppermost sheets from the stack of
sheets in tray 84 to a take away drive roll 90 and idler rolls 92.
Rolls 90, 92 guide the sheet onto transport 93 which in cooperation with idler
roll 95, de-skew rollers 71 and paper feed rollers 72 move the sheet to
transfer station D.
After transfer station D, photoconductive belt 10 passes beneath a corona
generating device 94 which charges any residual toner particles remaining on
belt 10 to a polarity conducive to their removal from photoconductive belt 10.
Thereafter, a pre charge erase lamp (not shown), located inside photoconductive
belt 10, discharges the photoconductive belt in preparation for the next
charging cycle. Residual particles are removed from belt 10 at cleaning station
G by an electrically biased cleaner brush 96 and two de-toning rolls 98 and
100.
It can be observed from the above description that the use of motors is
extensive throughout the xerographic process. Suitable motors or drive elements
are necessary for such operations as photoconductor belt movement, document to
platen movement and the general movement of copy sheets throughout the process
e.g., to the registration, transfer, fusing, and finisher stations.
Thus, there are ample opportunities for the use of improved motor techniques to
contribute to the efficiency, reliability, timing and synchronization of the
xerographic process, including distributed drive systems and the use of stepper
motors.
The self encoding technique described herein utilizes the fact that a
relationship exists between the back EMF of the motor coils and the shaft
position. The technique requires virtually no additional hardware for the
sensing of the coil back EMF. The back EMF is estimated by using the voltages
and currents of the two motor coils.
Note that the discussion that follows refers to a 2-phase, 200 step/rev,
bipolar hybrid stepping motor but that the same technique would be valid for
5-phase hybrid motors, or unipolar hybrid motors or 400 step/rev motors,
etc.
First, it will be shown how to analytically derive the shaft position of the
motor from the coil voltages and currents. The following are model equations,
in the Laplace domain, of the two motor coils:
where:
('1)
V_{COIL1} voltage in coil 1 [volts]
V_{COIL2} voltage in coil 2 [volts]
I_{COIL1} current in coil 1 [amps]
I_{COIL2} current in coil 2 [amps]
emf_{1} back emf of coil 1 [volts] which is equal to -K_{t}
ωsinθ
emf_{2} back emf of coil 2 [volts] which is equal to -K_{t}
ωcosθ
ω velocity of the motor shaft [shaft rad/sec]
θ position in the electrical cycle [electrical rad] (Note that for a 200
step/rev hybrid motor there are 50 electrical cycles per motor shaft cycle,
i.e., a full electrical cycle constitutes 4 full steps)
R coil resistance [amps]
coil inductance respectively [henries]
K_{t} motor back EMF constant (Volt-sec/rad)
Note: all variables are implicitly functions of time unless otherwise noted.
Specifically, a variable in the Laplace domain is denoted in standard form as a
function of s, the Laplace operator. It is assumed that intra-motor variations
in coil properties are minimal and therefore: R=R_{1} =R_{2}
and L=L_{1} =L_{2.} With this assumption equation (1') is
rewritten as
(1)
The two back-emf terms in equation (1), emf_{1} and emf_{2,}
are equal to K_{t} ωsinθ, K_{t} ωcosθ respectively in the
time domain. If it was possible to directly measure these voltages, the
electrical position, θ, could be found by the following:
(2)
Unfortunately, the back EMF cannot be directly measured without the use of
sensing coils. Additionally, as mentioned above, an op amp simulation of the
coil would require a differential component to simulate the voltage drop across
the inductor. A different approach must be taken to use the relationship
defined in equation (1).
After multiplying both sides of equation (1) by the quantity, 1/(sL/R +1) we
have:
(3)
In equation (3), the left hand side can be viewed as the two back EMF voltages
after being low passed filtered. Knowledge of I_{COIL1,}
I_{COIL2} and measurement of V_{COIL1} (s)/(sL/R+1) and
V_{COIL2} (s)/(sL/R+1) will make it possible to evaluate the left hand
components of equation (3) and use them to determine the electrical position,
θ, using a scheme like that shown in equation (2).
Many stepper motors used in industry have windings that are driven by current
chopping loops. The current chopper is a closed loop on current that controls a
switching amplifier (usually an H-bridge). The closed loop current control
regulates the coil current to some desired value by constantly switching the
coil voltage between +V_{S} and -V_{S} (where V_{S} is
the supply voltage).
The value of the: desired coil current is an input to the current chopper that
is commanded by the user, i.e., I_{COIL} is explicitly known because
the user generates the current command and therefore knows the current in each
of the motor windings.
The voltage in the motor coil is controlled by the current loop and not known
by the user, A measurement scheme must be employed to measure the coil voltage.
Direct measurement of the coil voltage is one alternative. Unfortunately,
direct measurement requires the use of a differential amplifier, since neither
side of the coil is referenced to a fixed voltage. The implementation of a
differential amplifier with switching inputs is cumbersome and, as will now be
shown, unnecessary.
View larger image here.
There are some properties of the full bridge amplifier, e.g., the SGS-Thomsen
L6203, as shown in FIG. 2, that allow clever indirect measurement of the coil
voltage. For simplification, it is assumed that the amplifier is ideal, i.e.,
there are no voltage drops across the transistors. At a later time, this
simplification will be removed but to ease the understanding of the development
that follows, an ideal amplifier is assumed.
Note than when the input voltage at the node IN1 (pin 5 in FIG. 2), hereafter
referred to V_{IN1,} is high (+5 V), the voltage to the motor coil is
+Vs. Similarly, when the input V_{IN1} is low (0 V), -Vs is the voltage
to the motor coil. Use these points to describe a straight line that estimates
V_{COIL,} given V_{IN1.} as:
or similarly
(4)
As stated, equation (4) is only valid when V_{IN1} is either zero (0)
or five (5) volts, since IN1 is a digital input. Is it important to realize,
however, that V_{IN1} is part of a closed loop design and has it's
value updated every 10 usec. Therefore, V_{IN1} contains frequencies
(up to 50 Khz) that are much faster than the electrical and mechanical time
constants of the motor and therefore do not affect the overall performance of
the system.
Additionally, the H bridge amplifier simply converts an input voltage level to
a corresponding output voltage level with no dynamics. This allows equation (4)
to be extended to estimate the mean or filtered value of V_{COIL} given
V_{IN1.} For example, if the mean of V_{IN1} is 3.0 volts, i.e.
assuming a 60% duty cycle, then the mean of V_{COIL} would be
V_{S} /5.
The use and analysis of PWM H bridge amplifiers in lieu of an analog amplifier
are common to those skilled in the art; their use is ubiquitous in cases such
as this where the effects of the higher frequency components of V_{IN1}
and hence V_{COIL} on the motor current are negligible due to the low
pass filter characteristics of the motor coil.
Extending (4), we have
or similarly
(4a)
where V_{COIL} and V_{IN1} are the filtered or mean values of
V_{COIL} and C_{IN1} respectively. Hereafter, terms with a
double underbar, e.g. V_{COIL,} represent low pass filtered or mean
values of the non-underlined variable.
Equations (4) and (4a) allow the determination of the coil voltage by observing
the logical input to the amplifier. In addition, if V_{IN1} is sent
through the RC filter shown in FIG. 2, we can use equation 4, and the Laplace
transform to define V_{F} (see FIG. 2) as follows:
(5)
After several time constants of R_{f} C_{f} have passed (steady
state), equation (5) can be approximated by:
(6)
or, alternately,
(7)
Recall now the form of the first term on the right hand side of equation (3),
i.e., V_{COIL1} (s)/(sL/R+1). If the R_{f} C_{f} time
constant in equation (7) is chosen to be equal to the L/R time constant seen in
equation (3), the two expressions will be equivalent. Thus, after choosing
R_{f} C_{f} =L/R, the right hand side of equation (7) can be
substituted into equation (3) to yield:
(8)
The time domain equivalents of emf_{1} (s)/(sL/R +1) and
emf_{2} (s)/(sL/R+1), called lemf_{1} and lemf_{2}
respectively, can be calculated by transforming equation (8) back into the time
domain as:
(8a)
View larger image here.
View larger image here.
Equation (8a) represents an algorithm for determining the back EMF voltages
that can readily be performed in real time by an Intel 8096 microcontroller.
The 8096 has an internal A/D converter that can accept up to 8 analog inputs
from the outside world. The analog input pins of the 8096 allow the voltages
V_{F} of both coils (V_{F1} and V_{F2)} as seen in FIG.
3A, FIG. 3B and FIG. 3C to be sent directly to the microcontroller.
In our implementation, the 8096 samples V_{F1} and V_{F2} and
internally calculates the discrete time values of low passed back EMF of the
coils. Using equation (2), the 8096 calculates the electrical position,
θ_{f,} using an algorithm as illustrated in FIG. 4 .
The calculation based upon observation of V_{F1} and V_{F2}
from the current chopping hardware is:
(9)
Thus, it is not necessary to reconstruct the back EMF signals from the low pass
estimates of equation (8a). We can indirectly utilize these low pass filtered
values to estimate the electrical position. The use of filtered values,
however, will induce a measurement error in our estimate of the electrical
position.
Consider the case of velocity regulation, i.e., a motor running at constant
speed: the electrical frequency, ω_{elec,} for 200 step/rev motor is
50 times the desired shaft frequency, ω_{DC.} A DC measurement error,
θ_{shift,} can be attributed to the DC phase shift introduced by the
back EMF filter.
The amount of phase lag the filtering introduces is equal to the phase lag
caused by the R_{f} C_{f} filter at a frequency of
ω_{elec.} Numerically,:
(10)
The nominal value of ω_{elec,} called ω_{nom,} is known and
is the frequency of the microstepping current commands generated by the 8096.
Using ω_{nom} the estimate of the electrical position becomes:
(11)
where θ_{f} is given by equation (9).
The estimation of the back EMF voltage can be improved by including the
non-ideal properties of the H-bridge driver, e.g., the SGS-Thomsen L6203 as
seen in FIG. 2, that have been neglected previously. The transistors and
diodes, previously assumed to have no voltage drop or resistance, do have a
nominal voltage drop and small signal equivalent resistance that are both a
function of the motor current.
In addition, the circuit parameters, i.e. the voltage losses and resistance,
are different in the "charge" state than in the recirculation state. These
characteristics, along with other non-ideal phenomenon, i.e., DC error in the
commanded current, can be partially corrected for by performing an
initialization measurement on the hardware similar to a calibration
setup/procedure.
A simple lumped parameter circuit, FIG. 5 will be used to model the
driver/motor circuit combination. The calibration test will be used to
determine the equivalent DC offset voltage, V_{OFF,} and AC resistance,
R_{AC,} of the model. The test will utilize the aforementioned
assumption that the two motor coils are well matched. This property allows one
set of current reference commands to determine both parameters.
The microcontroller initiates the test by commanding zero amps to one coil, and
one amp to the other coil, i.e., I_{COI1} =)and I_{COIL2} =1.
The microcontroller then waits one second for both the shaft position and coil
current to reach their steady state values. When the current is at steady state
(i.e. di/dt=0), the inductance has no effect and can be ignored.
Similarly, when the motor shaft has come to rest, the back EMF voltage is zero
(recall that the back EMF voltage amplitude is proportional to ω), and the
model at steady state can be reduced to the lumped parameter circuit shown in
FIG. 6. The two coils then satisfy the following set of equations:
The above set of equations can be used to determine the model parameters:
(12)
(13)
For purposes of modeling and analysis, the value of R_{AC} should be
used as a replacement for R, the motor resistance, since the overall resistance
of the driver/winding combination is what determines the circuits time
constant. In particular, R_{f} and C_{f} must be chosen such
that R_{f} *C_{f} =L/R_{AC.} Finally, V_{OFF}
and R_{AC} must be used in the main estimation algorithm, equation (8),
to fully include the non-ideal characteristics of the amplifier:
(14)
There has been shown a method for determining stepper motor shaft position that
requires virtually no additional hardware from that used to microstep the
motor. In addition, the algorithm software processing required is simple,
allowing it to be easily implemented and quickly executed in any of the
microcontrollers commercially available, e.g., the Intel 8096.
Claim Your: Useful
"Arduino Software Guide" Right Now... |
Click here for more project ideas.
The Essential Guide to the DHT22/11 humidity sensor including a library code guide for the Arduino. Learn how to determine Humidity, Dew point and Heat Index.
The Essential Guide to the MAX7219; What it is and how you can easily use one in any of your projects.
How to Easily Use the DS18B20 Maxim One-Wire thermometer with Arduino example code showing external or parasitic power modes.
New! Comments
Have your say about what you just read! Leave me a comment in the box below.